“For a dual-output flyback power supply that provides actual power at both outputs (5V 2A and 12V 3A, both of which can achieve ±5% adjustment), when the voltage reaches 12V, it will enter a zero-load state and cannot Adjust within the 5% limit. A linear regulator is a feasible solution, but it is still not an ideal solution due to its high price and reduced efficiency.
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1. Ferrite magnetic amplifier in flyback power supply
For a dual-output flyback power supply that provides actual power at both outputs (5V 2A and 12V 3A, both of which can achieve ±5% adjustment), when the voltage reaches 12V, it will enter a zero-load state and cannot Adjust within the 5% limit. A linear regulator is a feasible solution, but it is still not an ideal solution due to its high price and reduced efficiency.
Our proposed solution is to use a magnetic amplifier at the 12V output, even the flyback topology can be used. In order to reduce costs, it is recommended to use ferrite magnetic amplifiers. However, the control circuit of the ferrite magnetic amplifier is different from the control circuit of the traditional rectangular hysteresis loop material (high permeability material). The ferrite control circuit (D1 and Q1) can absorb current in order to maintain power at the output. The circuit has been fully tested. The transformer winding is designed for 5V and 13V output. The circuit can achieve 12V output ± 5% adjustment, and even can reach input power less than 1W (5V 300 mW and 12V zero load).
figure 1
2. Use the existing arc suppression circuit to provide overcurrent protection
Consider the 5V 2A and 12V 3A flyback power supplies. One of the key specifications of the power supply is to provide over power protection (OPP) for the 5V output when the 12V output reaches no load or when the load is extremely light. Both of these output ends have a voltage regulation requirement of ± 5%.
For the usual solution, the use of a sense resistor will reduce the cross-regulation performance, and the fuse is expensive. And now there are arc suppression circuits for over-voltage protection (OVP). This circuit can meet OPP and voltage stabilization requirements at the same time, and this function can be realized by using part of the arc suppression circuit.
It can be seen from Figure 2 that R1 and VR1 form a 12V output active dummy load, so that 12V voltage regulation can be achieved when the 12V output is lightly loaded. When the 5V output terminal is overloaded, the voltage on the 5V output terminal will drop. The dummy load will draw a lot of current. The voltage drop on R1 can be used to detect this large amount of current. Q1 turns on and triggers the OPP circuit.
figure 2
3. Active shunt regulator and dummy load
In the field of switching power supply products from online voltage AC to low voltage DC, flyback is currently the most popular topology. One of the main reasons for this is its unique cost-effectiveness, which can provide multiple output voltages by simply adding additional windings to the transformer secondary.
Usually, the feedback comes from the output that has the most stringent requirements on the output tolerance. This output then defines the number of turns per volt for all other secondary windings. Due to the leakage inductance effect, the output terminal cannot always obtain the required output voltage cross-regulation, especially when the given output terminal may have no load or the load is extremely light due to the full load of other output terminals.
You can use a post-stage regulator or dummy load to prevent the output voltage from rising in such situations. However, since post-stage regulators or dummy loads will cause increased costs and reduced efficiency, they are not sufficiently attractive, especially in recent years for the no-load and/or standby input power consumption in a variety of consumer applications. With more and more stringent regulations, this design began to be ignored. The active shunt regulator shown in Figure 3 can not only solve the voltage regulation problem, but also minimize the cost and efficiency impact.
Figure 3: Active shunt regulator for multiple output flyback converters.
The working method of this circuit is as follows: When the two output terminals are in the voltage regulation range, the resistor divider R14 and R13 will bias the transistor Q5, and then keep Q4 and Q1 in the off state. Under such working conditions, the current flowing through Q5 acts as a very small dummy load at the 5V output.
The standard difference between the 5V output terminal and the 3.3V output terminal is 1.7V. When the load requires additional current from the 3.3V output terminal, and the load current output from the 5V output terminal does not increase by the same amount, its output voltage will increase compared with the 3.3V output terminal voltage. Since the voltage difference exceeds approximately 100 mV, Q5 will be biased off, thereby turning on Q4 and Q1 and allowing current to flow from the 5V output to the 3.3V output. This current will reduce the voltage at the 5V output terminal, thereby reducing the voltage difference between the two output terminals.
The amount of current in Q1 is determined by the voltage difference between the two output terminals. Therefore, the circuit can keep both output terminals at voltage regulation without being affected by the load. Even in the worst case of full load at the 3.3V output terminal and no load at the 5V output terminal, the voltage regulation can still be maintained. Q5 and Q4 in the design can provide temperature compensation, because the VBE temperature changes in each transistor can cancel each other out. Diodes D8 and D9 are not necessary devices, but can be used to reduce power dissipation in Q1, eliminating the need to add a heat sink to the design.
This circuit only reacts to the relative difference between the two voltages, and basically does not work under full load and light load conditions. Since the shunt regulator is connected from the 5V output to the 3.3V output, compared with the grounded shunt regulator, the active dissipation of the circuit can be reduced by 66%. The result is to maintain high efficiency at full load, and maintain a low level of power consumption from light load to no load.
4. High voltage input switching power supply using StackFET
Industrial equipment that uses three-phase alternating current to work often requires an auxiliary power stage that can provide stable low-voltage direct current for analog and digital circuits. Examples of such applications include industrial drives, UPS systems, and energy meters.
The specifications of this type of power supply are much stricter than those required by standard off-the-shelf switches. Not only are the input voltages higher in these applications, but equipment designed for three-phase applications in industrial environments must also tolerate very wide fluctuations—including extended drop times, surges, and accidental loss of one or more phases. Moreover, the specified input voltage range of this type of auxiliary power supply can reach as wide as 57 VAC to 580 VAC.
Designing a switching power supply with such a wide range can be said to be a big challenge, mainly due to the high cost of the high-voltage MOSFET and the limitation of the dynamic range of the traditional PWM control loop. StackFET technology allows the use of less expensive low-voltage MOSFETs with a rated voltage of 600V in combination with integrated power controllers provided by Power Integrations, so that a simple and cheap switching power supply that can work within a wide input voltage range can be designed.
Figure 4: Three-phase input 3W switching power supply using StackFET technology.
The circuit works as follows: The input current of the circuit can come from a three-phase three-wire or four-wire system, or even a single-phase system. The three-phase rectifier is composed of diodes D1-D8. Resistors R1-R4 can provide inrush current limit. If fusible resistors are used, these resistors can be safely disconnected during faults without the need for a separate fuse. The pi filter is composed of C5, C6, C7, C8 and L1, which can filter and rectify DC voltage.
Resistors R13 and R15 are used to balance the voltage between the input filter capacitors.
When the MOSFET in the integrated switch (U1) is turned on, the source of Q1 will be pulled low, R6, R7, and R8 will provide gate current, and the junction capacitance of VR1 to VR3 will turn on Q1. Zener diode VR4 is used to limit the gate source voltage applied to Q1. When the MOSFET in U1 is turned off, the maximum drain voltage of U1 will be clamped by a 450 V clamp network composed of VR1, VR2, and VR3. This limits the drain voltage of U1 to close to 450 V.
Any additional voltage at the end of the winding connected to Q1 will be applied to Q1. This design can effectively distribute the total amount of rectified input DC voltage and flyback voltage between Q1 and U1. Resistor R9 is used to limit the high frequency oscillation during the switching period. Due to the leakage inductance during the flyback interval, the clamping network VR5, D9 and R10 are used to limit the peak voltage on the primary.
Output rectification is provided by D1. C2 is the output filter. L2 and C3 form a secondary filter to reduce the switching ripple at the output.
When the output voltage exceeds the total voltage drop of the optocoupler diode and VR6, VR6 will be turned on. The change of the output voltage will cause the current flowing through the optocoupler diode in U2 to change, thereby changing the current flowing through the transistor in U2B. When this current exceeds U1’s FB pin threshold current, the next cycle will be suppressed. Output voltage regulation can be achieved by controlling the number of enable and inhibit cycles. Once the switching cycle is turned on, the cycle will end when the current rises to the internal current limit of U1. R11 is used to limit the current flowing through the optocoupler during transient loads and to adjust the gain of the feedback loop. Resistor R12 is used to bias Zener diode VR6.
IC U1 (LNK 304) has built-in functions, so it can protect the circuit according to the disappearance of the feedback signal, the short circuit of the output terminal, and the overload. Since U1 is directly powered by its drain pin, there is no need to add an additional bias winding on the transformer. C4 is used to provide internal power supply decoupling.
5. Choosing a good rectifier diode can simplify the EMI filter circuit in the AC/DC converter and reduce its cost
This circuit can simplify the EMI filter circuit in the AC/DC converter and reduce its cost.
To make AC/DC power supplies comply with EMI standards, a large number of EMI filter devices, such as X capacitors and Y capacitors, are needed. The standard input circuit of AC/DC power supply includes a bridge rectifier to rectify the input voltage (usually 50-60 Hz). Since this is a low-frequency AC input voltage, standard diodes such as 1N400X series diodes can be used. Another reason is that these diodes are the cheapest.
These filter devices are used to reduce the EMI generated by the power supply in order to comply with the published EMI limits. However, since the measurement used to record EMI only starts at 150 kHz, and the AC line voltage frequency is only 50 or 60 Hz, the standard diode used in the bridge rectifier (see Figure 1) has a longer reverse recovery time. And it is usually not directly related to EMI generation.
However, the input filter circuit in the past sometimes included some capacitors connected in parallel with the bridge rectifier to suppress any high-frequency waveforms caused by the rectification of the low-frequency input voltage.
If fast recovery diodes are used in the bridge rectifier, these capacitors are unnecessary. When the voltage between these diodes starts to reverse, their recovery speed is very fast (see Figure 2). In this way, by reducing the subsequent high-frequency turn-off sudden change and EMI, the stray line inductance excitation in the AC input line can be reduced. Since two diodes can be turned on in each half cycle, only two of the four diodes need to be of the fast recovery type. Similarly, of the two diodes that conduct conduction every half cycle, only one of the diodes needs to have fast recovery characteristics.
Figure 6: A typical input stage of an SMPS using a bridge rectifier at the AC input.
Figure 7: The input voltage and current waveforms show the diode sudden change at the end of reverse recovery.
6. Use soft start to prohibit low-cost output to curb current spikes
To meet strict standby power consumption specifications, some multiple output power supplies are designed to disconnect the output connection when the standby signal is active.
Normally, this can be achieved by turning off the series bypass bipolar transistor (BJT) or MOSFET. For low current output, if the extra voltage drop of the transistor is fully considered when designing the power transformer, BJT can become a suitable substitute for MOSFET, and the cost is lower.
Figure 10 shows a simple BJT series bypass switch with a voltage of 12 V, an output current intensity of 100 mA, and a super capacitor (CLOAD). The transistor Q1 is a series bypass element, and the switch is controlled by Q2 according to the state of the standby signal. The value of resistor R1 is rated, so that it can ensure that Q1 has enough base current to work in a saturated state under minimum Beta and maximum output current. PI recommends adding an additional capacitor (Cnew) to adjust the transient current during turn-on. If Cnew is not added, Q1 enters the capacitive load quickly after it is turned on, and thus produces a larger current spike. In order to adjust the transient spike, the capacity of Q1 needs to be increased, which leads to an increase in cost.
Cnew used as an additional “Miller capacitor” for Q1 can eliminate current spikes. This extra capacitance can limit the dv/dt value of the collector of Q1. The smaller the dv/dt value, the smaller the charging current flowing into Cload. Specify the capacitance value for Cnew so that the ideal output dv/dt value of Q1 multiplied by the value of Cnew equals the current flowing into R1.
Formula 2
Figure 8: A simple soft-start circuit can inhibit the power output during standby and eliminate current spikes during turn-on. Therefore, a small transistor (Q1) can be used to keep costs low
The Links: LB043WQ1-TD05 LQ038Q7DR01
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